Ultra-wideband monopole antenna with U and L shaped slotted patch for applications in 5G and short distance wireless communications

Abstract

In this paper, a monopole antenna with Ultra-wideband (UWB) characteristics is presented for applications in 5G and short-distance wireless communications. It consists of a rectangular shaped patch having truncated corners and central notch with an extended stub at top edge. Two layers of slots are engraved in the middle of the patch within which the outer layer contains four L shaped slots and the inner layer contains two face to face vertically placed U-shaped slots. A partial ground is structured with staircase slots to attain ultra-wideband response with sufficient gain and efficiency. The proposed antenna provides 124% of impedance bandwidth (2.8 GHz–12 GHz) implemented on a low-cost FR4 substrate having the electrical dimension of 0.3𝜆 × 0.19𝜆 at the low cutoff frequency. The effect of the inclusion of slots within the patch is clearly observable as it increases gain and efficiency. The proposed antenna exhibits better response with a maximum measured gain of 6.5 dBi and efficiency of 85% within the bandwidth. Near constant time delay around 1 nS and good fidelity factor of 0.8316 represents that this slotted antenna can be used for undistorted reliable signal transfer. This antenna also provides symmetrical radiation patterns with less cross-polarization effects. Therefore, this monopole antenna can be an appropriate candidate for short-range communication applications like 5G, wireless local area networks, transfer data from biomedical sensors and imaging.

Keywords

Ultra-wideband defected ground structure monopole slotted patch 5G wireless communication

1. Introduction

Ultra-Wide Band (UWB) system which covers 3.1 GHz–10.6 GHz for commercial applications can be used for radar [1], LEO, MEO satellites [2], biomedical sensing applications [3], remote sensing [4], wireless body area networks [5], internet of things (IoT) [6], moreover in 5G communications [7]. Numerous approaches have been taken by the researchers to design an antenna for achieving high gain, bandwidth, efficiency, and better radiation pattern within a small size for these applications. Antenna size can be minimized by using substrates having high permittivity or small size radiating elements [8]. Inclusion of slots in the patch, using defected ground structure, adding parasitic elements have the effects of dimension miniaturization and performance improvement of antennas.

As an earlier work, N. Behdad et al. demonstrated a UWB antenna in 2005 where UWB operation is originated by coupling to antennas magnetically where the antennas are sectorial loop-shaped. The average radius of each antenna is 27.5 mm placed on a large ground plane of dimension 20 × 20 cm². This antenna covers ultra-wide bandwidth ranging from 1.7 GHz–14.5 GHz. This antenna shows a distorted impulse response due to asymmetry in its structure [9]. A novel UWB antenna is presented by N. Telzhensky et al. where the genetic algorithm is used to optimize the size of the antenna setting the optimization goal of the smooth transfer function, low reflection coefficient, and less group delay deviation to ensure high correlation between transmitting and receiving antenna [10]. Different types of patch antennas are also presented with various dimensions, gains, and applications in [11–18]. A UWB monopole antenna having an extra narrow band for Wi-Fi application is proposed in [19], where the overall size of the antenna is 27 × 33 mm². The maximum gain of this antenna is 3.9 dB keeps a room for further improvement. A very compact UWB antenna for microwave imaging is designed associating with double negative metamaterial [20]. The dimension of this antenna is very small (16 × 21 mm²), and the maximum gain is 5.16 dB at high frequency. At low and mid-frequency low average gain imposes the necessity of further improvements. A MIMO antenna having dimension (22 × 38 mm²) is presented in [21] where the complementary split-ring resonator is used to get an ultra-wideband response. In [22–24], metamaterial loaded multiband antenna have been presented. Metamaterial based circularly polarized (CP) antennas are represented in [25–27], which have narrow impedance bandwidth, poor gain, higher design complexity, larger size. Due to these limitations, these antennas are not suitable in applications where higher bandwidth, higher gain, and compact size are necessary. A compact tapered slot antenna with differential microstrip feed lines are presented in [28] having bandwidth 3 GHz–11 GHz, which is suitable for differential RF front end of compact UWB systems. However, the fabricated antenna shows an out-band notch at within 8 GHz to 10 GHz, and low gain in low and high frequencies imposes a limitation of this antenna. A compact ultra-wideband (UWB) diversity antenna with inverted-L stubs and complementary split-ring resonator (CSRR) on the ground plane is introduced in [29], which are applicable for MIMO applications. A new flexible ultra-wideband (UWB) antenna is presented for wearable applications in the 3.7–10.3 GHz band, which is realized using conductive fabric embedded into polydimethylsiloxane polymer and tolerant to body loading and physical deformation [30]. However, the average gain of this antenna is less than four dBi in free space, whereas in deformation, state average gain comes below 0 dBi. It shows average efficiency in free space near 50%, which decreases to 30% in the bent state. In [31], a folded fin structure is exercised to get UWB response. This antenna shows better efficiency (more than 80%) in the lower end of bandwidth, but as frequency increases, efficiency decreases linearly. Dual-Band Fork-Shaped Monopole Antenna for Bluetooth and UWB applications is designed by Mishra et al. having a dimension of 42 × 24 mm². This antenna shows average gain within the UWB range is 3.8 dBi [32]. A folded feed, the stacked patch antenna is introduced in [33] where high average gain is achieved compensated by higher dimension (80 × 80 mm²) of the antenna. In [34], a band notch antenna of size (30 × 31 mm²) is described with average gain 4 dBi, which shows a reject band within 5–6 GHz. Recently, a diamond-shaped slotted antenna has been presented in [35], having structure 26 × 30 mm².

In this paper, a monopole antenna with slotted rectangular patch is presented. This antenna contains a simple rectangular patch, which includes corner cuts at the upper top edge. At the centre of the top edge of the patch, there is a horizontal notch within which extended stub is included. Two layers of slots are etched in the middle of the patch where the outer layer contains L shaped slots and inner layer comprises with two face to face vertical U slots. The impact of these slots noticed as they help to enhance efficiency and gain of the antenna. Asymmetric dimensions of the slots are chosen to get increased gain, bandwidth, and efficiency. The motivation of this design is to get ultra-wideband characteristics keeping the antenna size small without compromising with antenna performance in terms of gain, efficiency, radiation pattern, group delay, and fidelity factor. The effect of various change of patch, ground configuration, patch and feed position have also been studied. The Ultra-wideband antenna characteristic provides high resolution and accurate localization of a target for microwave imaging [36]. Since UWB antenna provides broad frequency band (3.10–10.6 GHz) so information can be collected over a broadband frequency range in a single test, leading to the feasibility of fast UWB data acquisition and rapid microwave detection. Since the proposed antenna is small in size, provides the broadband frequency (2.8 GHz–12 GHz) with efficiency more than 80%, average gain 5.5 dBi for this reason, this antenna is an appropriate candidate for a microwave imaging and 5G applications. This antenna can also be applicable for short-distance wireless local area networks and transfer data from biomedical sensors. The subsequent parts of this paper are organized as antenna details are discussed in Section 2. Parametric analysis is performed in Section 3. The equivalent circuit of the antenna is represented in Section 4 whereas Section 5 includes the result and discussion of the designed antenna, and a conclusion is drawn in Section 6.

2. Antenna design and geometry analysis

The proposed antenna has been designed with a rectangular patch having the monopole configurations. The antenna geometry is constructed on an FR-4 substrate having permittivity (ϵ_r) of 4.3 and permeability (μ_r) of 1. The thickness (h) of the substrate is 1.6 mm with a copper layer of thickness 0.035 mm at both sides of it. In the design process, at first, a theoretical calculation has been accomplished to determine the possible dimensions of the patch and substrate of the proposed antenna. Since frequency ranges from 3.1 GHz to 10.6 GHz is considered as the ultrawideband, a sampling frequency of 3.5 GHz within this range is considered as the resonance frequency (f _r) to calculate initial length and width of the patch. The width of the patch can be determined using the following equation [37]: $\begin{eqnarray}\displaystyle W_{p}=\frac{C}{2f_{r}}\sqrt{{\displaystyle \frac{2}{{\varepsilon}_{r}+1}}}. & & \displaystyle\end{eqnarray}$ (1) Since the fringing effect is obvious in the antenna so, effective permittivity (ϵ_reff) and effective patch length (L _eff) can be calculated by using the equation (2) [37]. $\begin{eqnarray}\displaystyle \text{} & \text{} & \displaystyle {\varepsilon}_{\mathit{reff}}=\frac{{\varepsilon}_{r}+1}{2}+\frac{{\varepsilon}_{r}-1}{2}\left[1+12\frac{h}{W}\right]^{-0.5}\end{eqnarray}$ (2) $\begin{eqnarray}\displaystyle \text{} & \text{} & \displaystyle L_{\mathit{eff}}=\frac{c}{2f_{r}\sqrt{{\varepsilon}_{\mathit{reff }}}}.\end{eqnarray}$ (3)

Extension of the patch and exact length of it can be represented by using equation (4) & (5) $\begin{eqnarray}\displaystyle \text{} & \text{} & \displaystyle {\rm\Delta}L=h\times 0.412\frac{({\varepsilon}_{\mathit{reff}}+0.3)\left({\displaystyle \frac{W}{h}}+0.264\right)}{({\varepsilon}_{\mathit{reff}}-0.258)\left({\displaystyle \frac{W}{h}}+0.8\right)}\end{eqnarray}$ (4) $\begin{eqnarray}\displaystyle \text{} & \text{} & \displaystyle L_{p}=L_{\mathit{eff}}-2{\rm\Delta}L.\end{eqnarray}$ (5)

Calculation of substrate length and width $\begin{eqnarray}\displaystyle \text{} & \text{} & \displaystyle L_{s}=6h+L_{p}\end{eqnarray}$ (6) $\begin{eqnarray}\displaystyle \text{} & \text{} & \displaystyle W_{s}=6h+W_{p}.\end{eqnarray}$ (7)

By using the above-mentioned equations, length and width of the patch are around 26.3 mm and 20.2 mm whereas substrate length and width is around 29.8 mm and 35.9 mm. Considering these parameters, the design of the basic antenna is performed by using numerical simulation software CST microwave studio 2017. To obtain the desired bandwidth of UWB frequency range, high efficiency and gain of the antenna, the dimension of the substrate and patch is modified from the calculated values. Since the position of the patch plays a vital role in controlling the antenna performance, so to obtain a better response, the patch is positioned at 4 mm away from the left edge of the substrate by checking through several experiments. A feed line of 2.97 mm width is placed at 2.45 mm from the left edge of the patch to maintain 50 Ω characteristics impedance with ultrawideband performances. The design process of the antenna undergoes various modifications in the patch, which is shown in Fig. 1. At the top, a notch has engraved within which at the middle, and an extended stub is added. Notch and stub create an E shaped upper part. Then slots are engraved in the middle of the patch. Then two corners at the top are chamfered to get ultra-wideband response. The ground plane is also altered to the partial ground with slots to obtain better efficiency and gain within the desired bandwidth.

Fig. 1.

Evolution steps of the patch of proposed antenna (a) design 1 (b) design 2 (c) design 3 (d) design 4 and (e) proposed design.

At first, a flat rectangular patch, as shown in Fig. 1(a) with the proposed ground is checked through simulation. Since microstrip patch antenna acts like an LC resonant circuit [38], dual-band response with bandwidth from 3 GHz–4.88 GHz and 5.6 GHz–11.25 GHz is observed. It is also noticed that at low frequencies, a moderate amount of current flows through the two vertical edges of the patch. In the middle of the patch, a uniformly distributed lower current flows. As the frequency increases, current density also increases from the side edge towards the centre. To increase the current density at the upper edge of the patch a notch is created in the middle of the top of the patch, which is shown in Fig. 1(b). Due to this notch, more current flows at the bottom of notch and two sides of the patch, and thus current flow increases, which causes to increase inductance. This notch also causes to increase current at the centre of the patch at low frequencies compared to earlier. Due to increased current, the increased radiation efficiency is observed. A change in stopband with increased |S ₁₁| occurs in the frequency range from 4.87 to 5.66 GHz. Another impact of this notch, to shift the resonance frequencies towards the lower frequencies, as depicted in Fig. 2(a). The change in the resonance frequencies is due to increase capacitance and inductance caused by a notch.

Fig. 2.

(a) |S ₁₁| of slotted monopole antenna for ultra-wideband application for a different change of patch and (b) |S ₁₁| of the proposed antenna for full ground, partial and defected ground.

Now, this antenna undergoes further alternation by adding an extended stub from the centre of the notch, as shown in Fig. 1(c). Since an open-circuit stub offers capacitive and inductive effect around resonance frequency and it shows dual-band characteristics [39] two sharp resonances are observed at 8.9 GHz and 10.4 GHz as shown in Fig. 2(a). Higher current flow surrounding this stub causes an improved mid-frequency gain. In this state, current at the middle of the patch is less compared to four sides of the patch. As the frequency increases, the current density increases from side to the middle of the patch. From the literature, it is found that slots inside the patch change the courses of surface current flow, which eventually causes to decrease the size of the patch [40,41] or to increase band width [42,43]. To study the effect of the slot, the patch is further modified by inserting slots in the middle of the patch, as shown in Fig. 1(d). Due to the insertion of these slots in the patch, improved gain and radiation efficiency are observed at mid and high frequencies. Current follows a path near the slots, and the current path is increased, which ultimately increases inductive reactance as frequency increases [44]. As the frequency increases, the increased current density is observed around these slots, which causes to improve |S ₁₁| values at high frequencies and a sharp resonance occurs at 9.97 GHz. Improved efficiency and gain are the ultimate effects of the insertion of these internal slots, which is depicted in Figs 3(a)–(b). At this stage, a uniformly distributed lower current flows through both upper corners of the patch at low frequencies. Sharp bend at two upper corners may impose larger reflection within frequencies 4.64 GHz–5.56 GHz. Due to this reflection, there exists a stopband near 5 GHz, as shown in Fig. 2(a), which hinders obtaining ultra-wideband response. To avoid this limitation, two corners are chamfered, considering an angle 45° that is shown in Fig. 1(e). This modification causes surface current to increase through these corners and a simulated −10 dB bandwidth is observed throughout the frequency range 3 GHz–11.1 GHz at the expense of a little decreasing mid frequency gain. The diagram of surface current distribution for different changes in patch is not included here for brevity. This proposed design is also verified with full ground, which shows narrow bandwidth at very high frequencies. By using the partial ground, it exhibits a better |S ₁₁| response with a decreased realized gain but proposed slotted partial ground provides the desired |S ₁₁| response with improved gain and efficiency. The |S ₁₁| for the various ground is shown in Fig. 2(b). The final design layout of the patch and ground of the proposed monopole antenna is depicted in Fig. 4(a)–(b) respectively. Moreover, dimensions of the substrate, patch are depicted in Tables 1 and 2 whereas Table 3 exhibits grounds parameters.

Fig. 3.

(a) Simulated realized maximum gain for different design steps and (b) Simulated efficiency graph of different stages during the developing of the proposed antenna.

Fig. 4.

(a) Patch geometry and (b) internal slot geometry of patch (c) Ground geometry of the designed antenna.

Since the effect of inclusion of slots on monopole and the partial ground has an impact on bandwidth enhancement as presented in Fig. 2(a) and 2(b) respectively, to explain this effect a comparison of the surface current has been done for different ground structure. Figure 5 shows the surface current pattern at 3.4 GHz for a full ground plane. From this Figure, it is observed that for full ground plane current concentration is more near the feedline compared to rest part of the monopole and current is distributed all over the ground plane with lower magnitude. On the other hand, in the case of a partial backplane, high dense direction current is observed in the backplane as shown in Fig. 6. In the chiselled slot on monopole, current concentration is high near the slots which contributes to the higher radiation and less reflection. Thus it increases the bandwidth.

Table 1

Dimension substrate and patch of the proposed antenna

Parameter	Dimension (mm)	Parameter	Dimension (mm)	Parameter	Dimension (mm)
L	20.5	W	32.5	h	4
FL	6	D1	3.1	D1, D3	1, 1
FW	2.97	D4	0.75	h ₁	6.45

Table 2

Various parameters of internal slots of the patch

Parameter	Dimension (mm)	Parameter	Dimension (mm)	Parameter	Dimension (mm)
L1	12.25	W1	13.45	W7	4.5
L2	5	W2	3	D1	0.8
L3	3.2	W3	2.5	D2, D3	1, 1
L4	5.5	W4, W5	5, 5	D4	0.95
L5, L6	4.2, 4.2	W6	5	D5	1

Table 3

Ground dimension of the proposed antenna

Parameter	Dimension (mm)	Parameter	Dimension (mm)	Parameter	Dimension (mm)
GW1	2.5	GW2, GW3	3, 3	GW4	2.75
GW5	3.75	GW6	0.55	GW7	3
GW8, GW9	3.5, 3.5	GL1	4.75	GL2, GL3	3.75, 2.75

Fig. 5.

Surface current distribution at 3.4 GHz of the proposed patch with full ground.

Fig. 6.

Surface current distribution at 3.4 GHz of the patch with partial ground.

Figure 7 exhibits the surface current distribution at different resonance frequencies of the proposed antenna. In this Figure, a noticeable fact is that slotted partial ground has a high concentration of current thus it contributes to obtaining the high bandwidth. From surface current distribution, as depicted in Fig. 7(a), it is observed that at 3.4 GHz where first resonance occurs, evenly distributed lower current flows at all over the patch. The current density is a little bit higher at the edges of the patch compared to the centre. Due to the lower current density, gain and efficiency are also low in lower frequency. As the frequency increases, the current density increases gradually near the edges of patch and ground slots, and at 8.8 GHz a sharp resonance is observed in this frequency as shown in Fig. 7(b). High current density exists almost all over the patch at this frequency. Especially higher dense current at near the outer slots, the upper notch of the patch, and ground slots indicate their contribution to better gain and efficiency along with sharp resonance. As the frequency increases, current concentration near the inner slots also increases, and a high current density around the inner slot is observed at 10 GHz, which is shown in Fig. 7(c). At this state, current concentration is dominant around the middle slot of the ground. Further increasing of frequency causes current to decrease, which affects the gain and efficiency, and current density is prominent around the slots of the left side of the patch, which is shown in Fig. 7(d). All these observations represent the contribution of the internal slots to raise gain and efficiency at mid and high frequencies along with the desired ultrawide bandwidth.

Fig. 7.

Surface current distribution of proposed antenna (a) 3.4 GHz (b) 8.8 GHz (c) 10 GHz and (d) 10.8 GHz.

Electric field distribution has also examined to realize the ultawideband performance. Figure 8 electric field distribution on the chiseled slots on the monopole at different resonance frequencies. A close observation of this Figure reveals that high electric field is obtained around the slots at different frequencies. These high electric fields contribute for radiating electromagnetic energy with less reflection coefficcient for broad range of frequencies. Thus chiseled slots on the monopole helps to increase the bandwidth of the antenna.

Fig. 8.

Electric field distribution on monopole at different frequencies (a) 3.4 GHz, (b) 8.8 GHz, (c) 10 GHz and (d) 10.8 GHz.

3. Parametric analysis of proposed antenna

The performance of the proposed antenna is further analyzed by investigating |S ₁₁| response for different positions of the patch and feedline from the left edge of the substrate. As represented in Fig. 9(a), it is observed that the position of the patch plays an important role in achieving the desired ultra-wideband response. When h = 1 mm, the antenna displays a dual-band response having a narrow stopband near 6 GHz. A broadband response extended from 3 GHz to 9 GHz and a sharp narrow band near 10 GHz are observed when h = 2 mm, whereas when h = 3 mm, the bandwidth of this narrow band is extended. When h = 5 mm, multiband resonance property is observed. But at h = 4mm, the antenna exhibits the desired ultra-wideband response. The position of the feedline is also an important factor and appropriate position of it necessary to get an ultra-wideband response, which is clearly observable from Fig. 9(b). Depending on the position of the feedline, the value of patch inductance alters since the current path is modified. For this reason, a variation in resonances and bandwidth is exhibited by the antenna for different positions of the feedline. When the feedline is near the centre of the patch (h ₁ = 8.45 mm), the antenna shows a triple band characteristic with three sharp resonances near 3.5 GHz, 7.5 GHz, and 11.5 GHz. Now changing the position of feed line to a place where h ₁ = 7.45 mm, a dual-band response is observed with the upper band extended from 6.5 GHz to 11.5 GHz. If h ₁ = 5.45 mm, it also represents dual-band properties with a wide band extended from 3 GHz to 9 GHz, and a narrow band response is observed near 11 GHz. When feedline placed at 6.45 mm, the desired ultra-wideband response is witnessed. The effect of this feedline position is further observed by determining the axial ratio to know the polarization behaviour of the proposed antenna. Figure 10 represents the axial ratio plot with respect to the frequency. From the graph, it is noticed that in the UWB range (3.1 GHz–10.6 GHz) axial ratio is more than 3 dB indicating linear polarization characteristics of the proposed antenna in UWB frequency region.

Fig. 9.

|S ₁₁| of the proposed antenna (a) for different position of the patch and (b) for different position of feedline.

Fig. 10.

Axial ratio plot of the poposed antenna.

Fig. 11.

Equivalent circuit of the proposed antenna.

Fig. 12.

Comparison of |S ₁₁| obtained from the equivalent circuit by using ADS with |S ₁₁| obtained from CST.

4. Equivalent circuit of proposed antenna

Various approaches can be followed to draw an antenna equivalent circuit of the patch antenna. In accordance with the cavity model, a microstrip patch antenna can be represented by a parallel RLC circuit [45]. The lumped equivalent model of antenna considers microwave elements in terms of resistance, inductance, capacitance, and conductance [46]. In antenna, the conductor has the effect of resistance, R, and it is associated with inductance in accordance with the ampere’s law. Since patch and ground are separated by the substrate, which is a dielectric material, a parallel combination of capacitance and conductance can be assumed. In Fig. 11, an approximate circuit diagram proposed antenna is shown where R, L, C form a resonant circuit for a simple rectangular patch. In our design at the upper side of the patch, there are two notched slots, and within these slots, there is a stub. This portion is like a horizontal E shaped pattern that has the effect of increased series inductance with this resonance circuit [44] and increased capacitance [47]. R₁, L₁, C₁ are equivalent lumped elements for this portion of patch and combined connected to the RLC resonance circuit in series. There are two layers of internal slots in the proposed antenna. The outer layer has four slots, and equivalent circuit of this slots are represented by inductors L₂, L₃, L₆, L₇ and capacitors C₂, C₃, C₆, C₇ where two left side slots are considered by two series-connected LC resonant circuits, and similarly right sides slots form another series circuit containing two LC circuits. These two circuits are represented by two parallel branches at both sides of the previous resonance circuit. Two slots of the central layer have the same effect of having inductance and capacitance and these are represented by inductances L₄, L₅ and capacitances C₄, C₅ and they formed two parallel resonance circuits on both sides of center resonance circuit. All these branches are coupled two each other by using coupling capacitors CC₁, CC₂, CC₃, CC₄ and in this design, all the shunt components are neglected to make the circuit simple one. The equivalent circuit is validated by determining |S ₁₁| response using advanced design system (ADS) simulation tools. The component values are adjusted by tuning to get the desired ultrawideband response. The simulation output of ADS is compared with |S ₁₁| obtained from the CST and presented in Fig. 12. From this Figure, it is noticed that both the curves have a close similarity with each other thus it validates that the equivalent circuit represents our proposed monopole antenna. The component values are listed in Table 4.

Table 4
Component values of the equivalent circuit

Component Value Component Value Component Value Component Value

R₁ 62.5 Ω C₅ 0.21 pF CC₄ 0.61 pF L₅ 0.3 nH

R 26.7 Ω C₆ 0.72 pF C 0.54 pF L₆ 7.72 nH

C₁ 0.54 pF C₇ 0.44 pF L₁ 4.3 nH L₇ 0.67 nH

C₂ 0.81 pF CC₁ 2.9 pF L₂ 0.2 nH L 0.36 nH

C₃ 1.35 pF CC₂ 2.85 pF L₃ 0.81 nH

C₄ 0.3 pF CC₃ 3.0 pF L₄ 0.52 nH

Component	Value	Component	Value	Component	Value	Component	Value
R₁	62.5 Ω	C₅	0.21 pF	CC₄	0.61 pF	L₅	0.3 nH
R	26.7 Ω	C₆	0.72 pF	C	0.54 pF	L₆	7.72 nH
C₁	0.54 pF	C₇	0.44 pF	L₁	4.3 nH	L₇	0.67 nH
C₂	0.81 pF	CC₁	2.9 pF	L₂	0.2 nH	L	0.36 nH
C₃	1.35 pF	CC₂	2.85 pF	L₃	0.81 nH
C₄	0.3 pF	CC₃	3.0 pF	L₄	0.52 nH

Fig. 13.

Photographs of the fabricated antenna (a) front view and (b) back view.

5. Experimental results and discussion

CST Microwave studio is used for simulation of the proposed antenna. The performance of the antenna is further verified through cross-checked by using HFSS software. The proposed antenna is fabricated and prepared for measurement, which is shown in Fig. 13. |S ₁₁| measurement has been done by using a Performance Network Analyzer (PNA) (10 MHz–67 GHz) as per the arrangement shown in Fig. 14. The measured |S ₁₁| value is compared with the simulated data of the proposed antenna. As depicted in Fig. 16(a), measured data shows a bandwidth from 2.8 GHz–12 GHz, whereas simulated bandwidth is 3.1 GHz–11 GHz. From Fig. 16(a), it is also observed that lower resonance frequency is well-matched with a simulated one, and the measured reflection coefficient at first resonance is very less compared to simulation. The second resonance occurs at 7.2 GHz, whereas the proposed one is at 8.8 GHz. The shift of resonance may be caused by the soldering effect, which triggers to change the values of circuit elements. The third resonance of the measured antenna takes place in between the third and fourth of the proposed one.

Fig. 14.

Photograph of |S ₁₁| measurement using network analyzer.

Fig. 15.

Photograph of radiation pattern measurement using Satimo near field measurement system.

Fig. 16.

(a) Comparison of measured |S ₁₁| with simulation in HFSS and CST and (b) measured gain comparison of the proposed antenna with simulation in HFSS and CST.

Fig. 17.

Simulated and measurement radiation efficiency.

A Satimo Star-Lab near-field antenna measurement system has been utilized to measure radiation characteristics, realized gain, and efficiency. This measurement set up is represented in Fig. 15. Figure 16(b) shows a comparison of the antenna gains between measured and simulation. Measured antenna gain is well above the simulated gain. A lower gain (3 dBi) is observed between 3 GHz to 5 GHz, but from 5 to 8.5 GHz, it shows high gain around 6 dBi. After 10.5 GHz gain decreases gradually. In Fig. 16(b), a noticeable variation between simulated (CST and HFSS) and measured results is observed. It is worthy to mention that, within 5 to 7 GHz, there is a large mismatch between the simulated and measured result. In the Satimo Star-Lab near-field antenna measurement setup, measurement is taken in two steps. In the first step, the measured result is achieved from 1–6 GHz (through satimo antenna range from 700 MHz to 6 GHz). In the second stage, it is taken from 6–12 GHz (through satimo antenna range from 6 GHz to 18 GHz). There is some amount of calibration mismatching within these two ranges. Due to the addition of this calibration error with the measured result, a large variation in gain is noticed within 5–7 GHz. Rippling in the measured result is also noticed, it may be caused, due to noise effects during measurement. Besides, fabrication of the prototype and soldering may also some effect to have mismatch between simulated and measured results. The measured radiation efficiency as depicted in Fig. 17, shows a close agreement with the simulated efficiency. It also provides an average efficiency of nearly 85%. This high efficiency is obtained due to the insertion of chiselled slots within the patch. Due to the inclusion of these slots, the surface current is not spread all over the patch rather it concentrates near the narrow region as shown in Fig. 7. This concentrated current contributes to increasing the radiation efficiency, thus above 80% efficiency is observed in the operating band.

The 2D radiation patterns of E and H plane are depicted in Fig. 18 at frequencies for 3.1 GHz, 5.2 GHz, 10.1 GHz. The radiation patterns are normalized to zero scales concerning their maximum value. The Theta (θ) and Phi (φ) are spherical coordinates to the Cartesian axis alignments such as, when φ is equal to constant 0° then, θ is equal to 0° to 360° is called the XZ cut and also called E-plane of the radiation pattern. On the other hand, when φ is equal to constant 90°, then θ is equal to 0° to 360° is known as YZ cut, as well as it is also called as H-plane. It has been observed that the proposed prototype offers an omnidirectional steady radiation pattern in the operating band. However, it is observed from the surface current distribution that at the lower frequencies the current density is lower than the higher frequencies. On the other hand, the fixed amount of excitation power is distributed to all frequency bands. As a result, a sufficient amount of power is not accumulated to excite different modes efficiently. Therefore, due to these reasons, the antenna shows nearly omnidirectional radiation pattern and low gain at lower frequencies such as 3.1 GHz. At high frequencies such as 5.2 GHz and 10.1 GHz, high current density with proper current distribution is observed which helps to obtain high gain at these frequencies. The patch topology of the designed antenna is not symmetric. Due to this unsymmetrical patch cuts and slots, phase distribution becomes unequal. It affects the radiation pattern, and it is not entirely stable, and at some frequencies, cross-polarization dominates over the cross-polarization. A little disagreement is observed between simulated and measured patterns. It may be due to fabrication and measurement errors. The 3D radiation patterns for the sake of simplicity are also plotted in Fig. 19 at discrete frequencies i.e. 3.1 GHz, 5.2 GHz, 6.5 GHz and 10.1 GHz for the proposed antenna configuration. These radiation patterns are omnidirectional at the lower and middle frequencies. However, a little bit directional pattern is observed at a higher frequency band i.e. 6.5 and 10.1 GHz, respectively.

Fig. 18.

Radiation pattern of proposed antenna at different frequencies (a) 3.1 GHz (b) 5.2 GHz (c) 10.1 GHz.

Fig. 19.

Simulated 3D radiation pattern (a) 3.1 GHz, (b) 5.2 GHz (c) 6.5 GHz (d) 10.1 GHz.

In the case of a small UWB antenna, its shape has a noticeable impact on gain stability and pulse spreading throughout the bandwidth [48]. For this reason, time-domain performance measurement is very important for a pulse-based system such as radar and imaging applications. Group delay, which can be defined as a negative derivative of phase response with frequency [49], can be used for time-domain performance measurement. It indicates the time delay that is encountered by impulse signal in proportion to various wavelength dimensions of the antenna. The group delay of the proposed antenna is measured by setting two antennas face-to-face and side by side. The measured data are compared with the simulation result, which is depicted in Fig. 20(d). To measure group delay, two antennas are kept 24 cm away to avoid the far-field effect. Considering the speed of light, the ideal delay is about 0.8 ns. From measurement, it is observed that our group delay is nearly 1 ns. The difference is attributed due to the additional constant delay caused in the receiving antenna. The small ripples in group delay in the frequency domain are observed as illustrated in Fig. 19(d) since antenna acts as a resonator consisting of inductor and capacitor, phase of the signal is not constant rather, it changes with frequency. A dominant shift of the phase occurred around 9 GHz due to enhanced inductive effect for large current concentration near the slots, which is obvious in Fig. 7(b), and it causes a sharp peak in the group delay graph.

Time-domain performance is also observed by investing pulse spreading phenomena by applying a short impulse where the sending and receiving antenna are placed face to face, side by side, in the y-axis, and side by side in the x-axis. The input-output pulse response scenario is represented in Fig. 20(a)–(c). From the Figure, it is found that the received signal is in the same shape as the sending one; there is no distortion. Little spreading is observed when the antennas are in face to face and side by side in y-axis whether in case of side by side in x-axis spreading is more pronounced. To characterize the time-domain performance, the system fidelity factor (SFF) is the most used parameter, which is determined by the correlation between input and output pulse, and SFF is explained rigorously in [50]. Here fidelity factor is calculated by using Matlab code based on the equation presented in [51], $\begin{eqnarray}\displaystyle F=\max \frac{\displaystyle \int _{-\infty }^{\infty }m(t)n(t-{\tau})dt}{\sqrt{\displaystyle \int _{-\infty }^{\infty }|m(t)^{2}|dt\int _{-{\alpha}}^{{\alpha}}|n(t)^{2}|dt}} & & \displaystyle\end{eqnarray}$ (8) Where m (t), n (t) represents transmit and receive signals respectively and ${\tau}=\sqrt{LC}$ denotes the time delay between transmit and receive signals. Simulated fidelity factor for face to face, in the y-axis (FFY) and side by side in the y-axis (SSY) is 0.8021 and 0.8316 respectively whereas for side by side (SSX), the x-axis is 0.74. As fidelity factor of more than 0.8 is an indication of undistorted completely characterized received signal [52] so the proposed antenna can be reliably used for undistorted signal transmission and reception in the face to face and side by side, y-axis.

Fig. 20.

Normalized magnitude of input-output pulse where two antennas are positioned at 24 cm (a) side by side, X (b) side by side, Y (c) face to face, Y (d) group delay.

A comparison is made between the present work and some previous works, and it is represented in Table 5. The physical dimension, the electrical dimension of the antenna, bandwidth, bandwidth ratio, efficiency, gain, and fidelity factor are the parameters of this comparison. The electrical dimension is calculated at the lowest cut off frequency. From the comparison table, it is observed that [19] shows higher efficiency and bandwidth ratio compared to the proposed antenna but it exhibits less gain with exercising a higher dimension. Antenna presented in [29] shows better fidelity factor but lower gain and efficiency with higher dimension makes it lag related to the proposed one. Thus, proposed antenna shows its better gain, efficiency, bandwidth ratio within its moderate dimension compared to other state of arts presented in [27,28,30–34,51].

Table 5

Comparison between the proposed antenna and other works (antenna size, bandwidth, efficiency, gain, fidelity factor)

Paper	Dimension (mm²)	Electrical dimension	Bandwidth (GHz)	Bandwidth ratio	Efficiency (%)	Gain (dB_i)
[19]	27 × 33	0.29𝜆 × 0.33𝜆	3.2–14, 2.38–2.57	4.4:1	90	3.9
[19]	27 × 33	0.29𝜆 × 0.33𝜆	3.2–14, 2.38–2.57	4.4:1	90	3.9
[28]	23 × 30	0.23𝜆 × 0.3𝜆	2.95–11	3.72:1	-	4
[29]	23 × 29	0.23𝜆 × 0.29𝜆	3–12	4:1	82	4.7
[30]	80 × 67	0.99𝜆 × 0.83𝜆	3.7–10.3	2.8:1	55	4
[31]	40 × 55	0.51𝜆 × 0.62𝜆	3.8–11.6	3.1:1	80	-
[32]	42 × 24	0.43𝜆 × 0.25𝜆	3.1–10.6	3.4:1	-	3.8
[33]	80 × 80	1𝜆 × 1𝜆	3.75–9.86	2.63:1	-	6
[34]	30 × 31	0.31𝜆 × 0.32𝜆	3.1–10.6	3.42:1	-	4
[35]	26 × 30	0.27𝜆 × 0.31𝜆	3.09–12.75	4.1:1	-	-
[53]	60 × 60	0.53𝜆 × 0.53𝜆	2.67–13	4.86:1	-	4.2
Proposed	32.5 × 20.5	0.3𝜆 × 0.19𝜆	2.8–12.4	4.3:1	85	6.5

6. Conclusions

A U and L shape slotted patch compact monopole antenna is presented in this article for ultra-wideband applications. Different cuts in the upper edges of the patch together with defected ground structure provides a wide bandwidth from 2.8 GHz to 12 GHz within a dimension of 32.5 × 20.5 mm². Effect of the U and L slots is investigated by insertion of those in the middle of the patch, which contributes to the high gain of 6.5 dBi (maximum). The equivalent circuit of the proposed antenna is also analyzed. The antenna is fabricated, and the measured result is compared with simulation. The measured reflection coefficient, gain and efficiency are compared with the simulation and both are well-matched. It shows a nearly omnidirectional radiation pattern, near-constant group delay around 1 nS, and good fidelity factor of 0.8021, 0.8316 and 0.74, respectively. With the comparison of some existing works, it is found that the proposed antenna is better in terms of size and gain. This antenna is a good candidate for short-ranged ultra-wideband applications such as wireless local area networks, transfer data from biomedical sensors, imaging and 5G communication applications.

Footnotes

Conflicts of interest

The authors declare no conflict of interest.

Funding

This work is funded by the Deanship of Scientific Research (DSR) at King Abdulaziz University, Jeddah under grant No (RG-2-135-41).

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